Multi-channel tuner using a discrete cosine transform

ABSTRACT

A satellite receiver includes a multi-channel tuner for processing a plurality of different transponder signals to simultaneously provide a plurality of different bit streams from at least two of the transponder signals. The multi-channel tuner includes (a) a demultiplexer for demultiplexing a received signal representing the plurality of transponder signals into a plurality of sample signals, each transponder signal conveying a bit stream, (b) a plurality of bifurcated filters operative on the plurality of sample signals for providing a plurality of filtered signals, and (c) a discrete cosine transform element operative on the plurality of filtered signals for simultaneously providing at least two of the bitstreams.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is related to the copending, commonly assigned U.S.patent application Ser. No. 10/428,973, entitled “A Transform-BasedAlias Cancellation Multi-Channel Tuner” filed on even date herewith.

BACKGROUND OF THE INVENTION

The present invention generally relates to signal receiving devices, andmore particularly, to a multi-channel satellite signal receiver.

A conventional satellite receiving device, such as a direct broadcastsatellite (DBS) receiver, can tune to any one of a number of satellitetransponders, each transponder transmitting a downlink signal in aparticular frequency band. The transponder downlink signal typicallyrepresents a bit stream in a packet format, the packets conveying data,such as audio, video, programming information, etc., associated with oneor more broadcast channels or services. In this regard, each transponderis typically associated with a different set of broadcast channels. Assuch, a desired sports program may be found on one of the broadcastchannels associated with one transponder while a movie may be found onone of the broadcast channels associated with a different transponder.

Unfortunately, as noted above, such a conventional satellite receivingdevice only tunes to one downlink signal from one transponder at a time.This leads to a number of problems. For example, “channel surfing,”i.e., switching from one broadcast channel to another, may entailswitching transponders, which causes additional processing delays—delaysthat slow down the channel surfing process. Further, in households thatdesire to simultaneously watch, or listen, to programs associated withdifferent transponders—those households must spend more money topurchase, or lease, multiple conventional satellite receiving devices.

SUMMARY OF THE INVENTION

Therefore, and in accordance with the principles of the invention, areceiving device includes a multi-channel tuner for simultaneouslyprocessing a plurality of received signals, each received signalcorresponding to a bit stream. The receiver includes a receiver sectionfor providing a signal having a plurality of different frequencychannels, each frequency channel conveying a different bit stream and amulti-channel signal tuner operative on the signal for recovering thedifferent bit streams from at least two of the plurality of differentfrequency channels and for simultaneously providing the recovereddifferent bits streams, wherein the multi-channel signal tuner utilizesa discrete cosine transformation (DCT).

In one embodiment of the invention, the receiving device is a satellitereceiver. The satellite receiver comprises a multi-channel tuner thatincludes (a) a demultiplexer for demultiplexing a received signalrepresenting a plurality of transponder signals into a plurality ofsample signals, each transponder signal conveying a bit stream, (b) aplurality of bifurcated filters operative on the plurality of samplesignals for providing a plurality of filtered signals, and (c) adiscrete cosine transform element operative on the plurality of filteredsignals for simultaneously providing signals representing at least twoof the bitstreams.

In another embodiment of the invention, an integrated circuit includes atransform element for receiving a plurality of signals. The transformelement is operative on the received plurality of signals using adiscrete cosine transform to simultaneously provide signals representingat least two bit streams, each bit stream associated with a differenttransmission frequency band. Illustratively, each frequency band isassociated with a different transponder of a satellite cabledistribution network.

In another embodiment of the invention, the receiving device is asatellite receiver. The satellite receiver performs a multi-channeltuning method that includes (a) demultiplexing a received signalrepresenting a plurality of transponder signals into a plurality ofsignals, each transponder signal conveying a bit stream, (b) filteringthe plurality of signals to provide a plurality of filtered signals, and(c) transforming the plurality of filtered signals in accordance with adiscrete cosine transform for simultaneously providing signalsrepresenting at least two of the bitstreams.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates an embodiment of a multi-channel tuner;

FIG. 2 shows an illustrative frequency spectrum for a signalrepresenting 16 transponder channels;

FIG. 3 shows an illustrative filter for use in the multi-channel tunerof FIG. 1;

FIG. 4 illustrates the relationship between a carrier set and a type 4discrete carrier transform in accordance with the principles of theinvention;

FIG. 5 is an illustrative block-level diagram of a receiver embodyingthe principles of the invention;

FIG. 6 is an illustrative embodiment of a multi-channel tuner inaccordance with the principles of the invention;

FIG. 7 shows an illustrative embodiment of a bifurcated filter inaccordance with the principles of the invention;

FIG. 8 shows an illustrative bifurcated filter bank in accordance withthe principles of the invention;

FIGS. 9–16 show illustrative matrix values for use in transform element330 of FIG. 6;

FIG. 17 is an illustrative block diagram of a demodulator for use in theembodiment of FIG. 6; and

FIG. 18 is another illustrative embodiment in accordance with theprinciples of the invention.

DETAILED DESCRIPTION

Other than the inventive concept, the elements shown in the figures arewell known and will not be described in detail. Also, familiarity withsatellite-based program distribution is assumed and is not described indetail herein. For example, other than the inventive concept, satellitetransponders, downlink signals, a radio-frequency (rf) front-end, orreceiver section, such as a low noise block, and formatting and encodingmethods (such as Moving Picture Expert Group (MPEG)-2 Systems Standard(ISO/IEC 13818-1)) for generating transport bit streams are well-knownand not described herein. In addition, the inventive concept may beimplemented using conventional programming techniques, which, as such,will not be described herein. Finally, like-numbers on the figuresrepresent similar elements.

An embodiment of a multi-channel receiver 100 is illustrated in FIG. 1.Receiver 100 includes a low noise block (LNB) 205, an analog-to-digital(A/D) converter 210, and a bank of tuning elements 140-1 through 140-N.One or more satellites (not shown) transmit a plurality of downlinkradio frequency (RF) signals 201 in different frequency bands (orfrequency channels) associated with different transponders at the samepolarization. Each transponder-specific RF signal represents a differenttransport bit stream encoded, e.g., in accordance with theabove-mentioned MPEG2. The RF signals 201 may, e.g., be in the frequencyrange of 17 GHz (giga-hertz). Illustratively, RF signals 201 includes Nadjacent frequency channels, whose center frequencies are F₀ to F_(N−1),respectively. The channel spacing, F_(S), is illustratively uniform andequal to the separation between adjacent center frequencies, e.g.,F_(S)=F₂−F₁. As such, the total bandwidth of all frequency bands,F_(total), equals NF_(S). Each frequency channel contains a modulationon its center frequency (carrier) of bandwidth F_(bw) and has an excessbandwidth of x % and a guard band F_(gb), whereF_(gb)=(F_(s)−(F_(bw)[(100+x)/100])). For purposes of illustration, itis assumed that N=16, and F_(S)=29.164 MHz, which is also illustrativeof a sixteen transponder digital satellite system (DSS).

The RF signals 201 are received by one or more antennas (not shown) ofreceiver 100 for application to low noise block (LNB) 205. The latterdown shifts and filters the received RF signals 201 and provides asignal 206, which is a near base-band signal having a total bandwidthacross all channels of F_(total). For example, the lowest frequencychannel (e.g., channel 0) has a carrier F₀=F_(S)/2. This is furtherillustrated in FIG. 2, which shows the spectrum of the near base-bandsignal 206 for the 16 DSS channels. Returning to FIG. 1, signal 206 isconverted from the analog domain to the digital domain via A/D converter210, which samples signal 206 at a sampling rate, F_(samp), equal to orgreater than the Nyquist rate for signal 206. Illustratively,F_(samp)=2F_(total), i.e., the sampling rate is twice the totalbandwidth across all frequency channels, i.e., F_(samp)=2NF_(S). In thisexample, F_(samp)=933.12 MHz. A/D 210 provides a signal 214, which is adiscrete time sequence of samples representing the plurality oftransponder channels at the sample rate, F_(samp).

Signal 214 is applied to the bank of tuning elements 140-1 to 140-N.Each tuning element filters signal 214 at a particular one of the Ntransponder channels to simultaneously provide a respectiveinphase/quadrature (IQ) baseband signal representing the associatedtransport bit stream. For example, tuning element 140-1 includes abandpass filter 145-1, a decimate by N element 150-1 and a demodulator155-1. Bandpass filter 145-1 has a passband centered by F₀, and a stopband that attenuates the remaining transponder channels. As such,bandpass filter 145-1 filters signal 214 to provide a filtered signal146-1 that represents only channel 0 (Ch0). An illustrative blockdiagram of a representative filter 145, which provides an output signal146, is shown in FIG. 3. Filter 145, e.g., illustratively includes 256tap coefficients as represented by tap coefficient 165, delay elementsZ⁻¹ as represented by delay element 160, each of which provide a timedelay equal to ½(NF_(S)), and adders as represented by adder 170. Thesample data impulse response of bandpass filter 145 is the sampleimpulse data response of a low-pass filter (not shown) modulated by asample data representation of a cosine wave at the carrier frequency ofthe transponder channel to be received. The phase of this cosine wave isso arranged that zero degrees aligns with the center of the linear phaselow pass sample data impulse response. Since signal 146 is a sequence ofsamples that only represents those samples associated with a particulartransponder channel, this signal is now over-sampled at F_(samp). Thismeans that relative to a single received transponder channel the samplerate is much greater than required. As such, and returning to FIG. 1,signal 146-1 is now decimated by N (where N, again, is the number oftransponder channels) via element 150-1 to provide a sequence of samplesat 1/Nth the sample rate, i.e., 2F_(S), to demodulator 155-1. The latterincludes a half-channel spacing sine/cosine generator for demodulationat the associated transponder frequency, e.g., F₀. Demodulator 155-1provides an inphase/quadrature baseband signal representing theassociated transport bit stream.

It can be observed from FIG. 1, that N tuning elements are required tosimultaneously provide N transport bit streams. In addition, thefiltering process of each bandpass filter 145, must run at a very highclock rate. For example, the symbol rate of signal 214 is close to 1 GHz(giga-hertz). However, and in accordance with the principles of theinvention, it is possible to transform the architecture of receiver 100into an architecture such that the high speed pipelined calculationsfollowed by a decimation operation as illustrated in receiver 100 arereplaced with a number of lower speed parallel calculations, which arethen summed together.

In particular, assume that N=16 DSS transponder channels as illustratedin FIG. 2. Illustratively, A/D 210 samples signal 206 (describedearlier) at the Nyquist rate equal to 2NF_(S). Each transponder channelcarrier frequency, F_(ch), is located in the middle of the correspondingfrequency channel. As such, for N channels, numbered 0 to N−1:

$\begin{matrix}{{F_{C\; H} = {{c\;{h \cdot F_{S}}} + \frac{F_{S}}{2}}},} & (1)\end{matrix}$where 0≦ch≦N−1. The transponder carrier channel frequencies can benormalized to the sampling rate 2NF_(S). In this case, equation (1)becomes:

$\begin{matrix}{{F_{CHN} = \frac{\left( {{{2 \cdot c}\; h} + 1} \right)}{4 \cdot N}},} & (2)\end{matrix}$where F_(CHN) represents the normalized transponder carrier frequencyfor a particular channel, ch, where, again, 0≦ch≦N−1. The set ofnormalized transponder carrier frequencies is referred to as a carrierset. For N=16 transponder channels, there are 16 normalized transpondercarrier frequencies in the carrier set. In accordance with an aspect ofthe invention, I have observed that this carrier set corresponds to avariant of the type 4 Discrete Cosine Transform. Application of the type4 DCT in this application for complex modulations has an advantage inthat the IQ (in-phase/quadrature) modulation is maintained as a realnear base band signal for down stream conventional near base banddemodulation.

The equation for a N point type 4 (or IV) DCT is shown below:

$\begin{matrix}{{{{DCT}_{N}^{IV}\left( {i,j} \right)} = {\sqrt{\frac{2}{N}} \cdot {\cos\left( {2\;{\pi \cdot \frac{{2j} + 1}{4N} \cdot \frac{{2\; i} + 1}{2N}}} \right)}}},} & (3)\end{matrix}$where i is the time index and 0≦i≦N−1; and j is the frequency index(frequency channel number) and 0≦j≦N−1, and N is the number of frequencychannels. The relationship between the type 4 DCT and the carrier set isillustrated in FIG. 4 for N=16. In FIG. 4, it is assumed that the A/Dconverter (e.g., A/D 210) samples at 32 times the channel spacing sothat 16 channels are equally spaced between 0 and the Nyquist foldingfrequency. Continuous Cosines at F_(ch) of each frequency channel (Ch0to Ch15) are illustrated in FIG. 4. A sample data representation of eachcosine carrier at the A/D sample rate is represented by the black dotson the continuous cosines shown in FIG. 4. Along the axis labeled “Timein A/D Samples”, the relationship between the carrier set and the type 4DCT is noted. In the first 16 samples (1–16), each carrier is exactlyequal to a 16 point type 4 DCT, as signified by the axis label“DCT_(IV).” In the next set of 16 samples (17–32), each carrier is minusthe time flipped 16 point type 4 DCT, as signified by the axis label“−Time Flipped DCT_(IV).” The third set of samples (33–48) is exactlyequal to minus the type 4 DCT, as signified by the axis label“−DCT_(IV).” The final set of 16 samples (49–64) is exactly equal to thetime flipped type 4 DCT, as signified by the axis label “Time FlippedDCT_(IV).” The aforementioned pattern of sign changes and time orderflips continuously repeats for any desired length of carrier set. Thisgeneralizes to any number of channels N and any order DCT. Another wayto look at FIG. 4 is to view the axis labels as identifying theoperations required to be performed on a cosine carrier for the cosinecarrier to match a type 4 DCT. For example, for time samples 49–64, thecosine carrier must be “time flipped” to match a 16 point type 4 DCT.Therefore, and in accordance with another aspect of the invention, theearlier described filter 145 can be used as the basis for exploiting thesymmetries of a type 4 Discrete Cosine Transform.

Now, assume a single low pass finite impulse response (FIR) filter forprocessing the multi-channel transponder signal. Let a sample dataimpulse response (in the z-domain) of the FIR filter be:

$\begin{matrix}{{{H_{LPF}(z)} = {\sum\limits_{I = 0}^{{k \cdot N} - 1}\;{A_{I} \cdot z^{- i}}}},} & (4)\end{matrix}$where A₁ are the filter tap coefficients, z^(−i) are delay elements, kis the number of filter taps, 0≦1≦N−1, and N is the number of frequencychannels. In accordance with an aspect of the invention, thearchitectural trick for using a DCT is to divide each filter such thatdifferent subsets of the taps are separately accumulated to providepartial results that match the symmetry of a DCT. In this example, thedivision of the filter is a bifurcation such that the odd and even tapcoefficients are separately accumulated thus providing partial resultsthat match the symmetry of the DCT type 4 as illustrated in FIG. 4. (Itshould be noted that the inventive concept is not limited to abifurcation.) In addition, the tap coefficients, A₁, will be distributedamong different ones of the bifurcated filters.

Turning now to FIG. 5, an illustrative receiver 200 in accordance withthe principles of the invention is shown. Receiver 200 includes a lownoise block (LNB) 205, an analog-to-digital (A/D) converter 210, amulti-channel tuner 215 and a broadcast channel distributor 240. Asdescribed above, one or more satellites (not shown) transmit a pluralityof downlink radio frequency (RF) signals 201 in different frequencybands (or frequency channels) associated with different transponders atthe same polarization. RF signals 201 are processed by LNB 205 toprovide signal 206, which is a near base-band signal having a totalbandwidth across all channels of F_(total). (Again, this is illustratedin FIG. 2 for 16 DSS channels.) Signal 206 is converted from the analogdomain to the digital domain via A/D converter 210, which samples signal206 at a sampling rate, F_(samp)=2NF_(S), to provide signal 214, whichis a discrete time sequence of samples representing the plurality oftransponder channels. Signal 214 is applied to multi-channel tuner 215(described below), which, in accordance with the principles of theinvention, processes signal 214 to provide a number of simultaneous bitstreams from two or more transponder channels as represented by bitstreams 231-1 through 231-L, where 1<L≦N. It should be noted that thesesimultaneous bit streams are applied to broadcast channel distributor240, which processes each of the bit streams to provide data associatedwith virtual channels 240-1 through 240-K, where K>1. For example,broadcast channel distributor 240 decodes each of the bit streamsencoded, e.g., in accordance with the earlier-mentioned MPEG-2 SystemsStandard ISO/IEC 13818-1. As such, each of these virtual channelsrepresents content and/or services, for example, audio, video (e.g., aselected movie), electronic programming guide etc. It should be realizedthe although shown as separate signals 240-1 through 240-K, one, ormore, of these signals may be multiplexed together for transmission on abroadcast medium, e.g., a cable, or via wireless (such as Wi-Fi(Wireless Fidelity)). For simplicity, other input signals to broadcastchannel distributor 240 specifying selection of content and/or serviceshas not been shown. Likewise, other circuitry for delivering thecontent/services, which may, or may not, be a part of receiver 200 alsonot been shown.

An illustrative embodiment of a multi-channel tuner 215 in accordancewith the principles of the invention is shown in FIG. 6. Multi-channeltuner 215 includes demultiplexer (demux) 220, bifurcated filter bank325, transform element 330 and demodulators 335-1 to 335-L. Demux 220samples signal 214 at a demultiplexer sampling rate, F_(F), (orpost-decimation sampling rate) to provide a number of decimated samplestreams, 221-1 to 221-N, where N is the number of transponder channels,to bifurcated filter bank 325 (described further below). As such,F_(F)=2F_(S). The bifurcated filter bank 325 forms a filter input vectorto transform element 330 (described below), which recovers therefrom,via demodulators 335, a number of simultaneous bit streams from two ormore transponder channels as represented by bit streams 231-1 through231-L, where 1<L≦N.

In this illustrative embodiment, transform element 330 uses a type 4Discrete Cosine Transform (DCT) for processing the filter input vector.The output vector from transform element 330 is further processed by arespective demodulator, 335-i, at each of the transponder carrierfrequencies.

Bifurcated filter bank 325 includes a number of bifurcated filters,325-i, where 1≦i≦N, i.e., one bifurcated filter for a respective one ofthe N decimated streams from demux 220. Since all the bifurcated filtershave a similar structure, only one bifurcated filter is described indetail herein. An illustrative bifurcated filter 325-1 in accordancewith the principles of the invention is shown in FIG. 7. Bifurcatedfilter 325-1 receives decimated stream 221-1 from demux 220 and providestwo output signals: signal 506, also designated as the even outputsignal (E), and signal 507, also designated as the odd output signal(O). Bifurcated filter 325-1 includes a number of delay elements asrepresented by delay element 515, a number of coefficient multipliers asrepresented by coefficient multiplier 520 (also referred to as tapcoefficient 520) having a coefficient value of A₀, and two summingnetworks 505 and 510. The time delay elements, e.g., time delay element515, delay the signal by an amount of time equal to the inverse of thedemux sampling rate, i.e., 1/F_(F). As can be observed from FIG. 7,summing network 505 adds together those values from the even coefficientmultipliers to form the even output signal. Similarly, summing network510 adds together those values from the odd coefficient multipliers toform the odd output signal. It can be further observed that each summingnetwork 505 and 510 alternates the sign of the value from alternatecoefficient multipliers. For example, the value from the coefficientmultiplier 525 is inverted by summing network 505. In other words, thecoefficients for the even output signal are the even numbereddecimations by 2N alternated in sign, while the coefficients for the oddoutput signal are the odd numbered decimations by 2N. This alternatingin sign implements the required inversions in sign required at points intime as illustrated by the axis of the graph in FIG. 4 for matching atype 4 DCT. This bifurcating filter structure shown in FIG. 7 results inthe even tap partial sum being modulated by a direct type 4 DCT, whilethe odd tap partial sum is always modulated by a time reversed type 4DCT. This can be realized with a single N point type 4 DCT by summingthe vector of even taps partial sum to the vector of odd taps partialsum in reverse time order. In addition, it should be observed that inthis example there are k=N tap coefficients per bifurcated filter, i.e.,k=16, with respect to equation (4). As such, the total number of tapcoefficients in this example is 256, which are spread among the 16bifurcated filters as indicated in FIG. 7. In particular, bifurcatedfilter 325-1 illustratively includes the 16 tap coefficients: A₀, A₁₆,A₃₂, A₄₈, A₆₄, . . . , A₂₀₈, A₂₂₄, and A₂₄₀, arranged as shown.Similarly, bifurcated filter 325-2 (not shown) includes the 16 tapcoefficients: A₁, A₁₇, A₃₃, A₄₉, A₆₅, . . . , A₂₀₉, A₂₂₅, and A₂₄₁. Thispattern continues, such that the last bifurcated filter 325-N (notshown) includes the 16 tap coefficients: A₁₅, A₃₁, A₄₇, A₆₃, A₇₉, . . ., A₂₂₃, A₂₃₉, and A₂₅₅. (Referring briefly back to FIG. 3, it could beobserved that the taps of filter 145 are divided among the N bifurcatedfilters.)

It should also be noted that the effective N, or, e.g., 16, bifurcatedfilter responses have the same delay such that transform element 330processes N decimated samples at the same time. In other words, at aparticular time, t_(P), a filter input vector is formed for applicationto transform element 330. This filter input vector includes one samplefrom each of the decimated sample streams at a particular sampling time.Turning back briefly to FIG. 6, the filter input vector comprisessamples F1 through FN.

An illustrative bifurcated filter bank is shown in FIG. 8. The latterillustrates the combination of the even signals and odd signals fromeach of the N bifurcated filters via N adders 345-i, 1≦I≦N, to completethe match to the symmetry of a DCT. For example, the even signal frombifurcated filter 325-1 is combined with the odd signal from bifurcatedfilter 325-N via adder 345-1. This provides the above-noted requiredtime reversal illustrated in FIG. 4. As a result, bifurcated filter bank325 provides the filter input vector [F1, F2, . . . , FN] to transformelement 330 as illustrated in FIG. 6.

Having matched the symmetry of a type 4 DCT with the use of theabove-described bifurcated filter bank 325, transform element 330provides a DCT_(IV) transformation of the filter input vector, where:O _(C) =DCT _(IV) F,  (5)where F is the filter input vector, O_(C) is an output vector, theelements of which represents each of the N transponder channels, andDCT_(IV) is an implantation of an N-point type DCT. One illustrativeimplementation of transform element 330 is simply the implementation ofequation (3), shown above. Indeed, in accordance with an aspect of theinvention, any of the known algorithms to realize a type 4 DCT may beused, each having different computational efficiencies. However, for thehigh sampling rate applications targeted, a sparse factoring of the DCTIV matrix by exploiting relationships to the Discrete Fourier Transform(DFT) is preferred. One illustrative sparse factoring of an N=16 pointDFT IV is provided below. In particular, DCT_(IV) is:DCT _(IV) =CF·CE·CDFT0·CDFT1·CDFT2·CDFT3·CDFT4·CDFT5·CD·CC.  (6)This factorization requires 62 additions and 46 multiplications. If all16 transponder channels are simultaneously received this corresponds to3.875 adds and 2.875 multiplies per channel per near base band sample(the demultiplexer sample rate is 2F_(S) in Hz)). The matrices shown inequation (6) are illustrated in FIGS. 9–16. It should be noted that thenotation C(x) or S(x) illustrated in these figures represents theoperations cosine(x) and sine(x), respectively. Further, where clearlyidentifiable portions of a matrix are equal to zero, a single “0” isentered on that portion of the matrix, as illustrated in matrix CD ofFIG. 10.

As noted above, an advantage of the use of a type 4 DCT is that theelements of the output vector represent each of the transponder channelswherein the IQ modulation is maintained as a real near base band signalfor down stream conventional near base band demodulation. As such, eachelement of the output vector from transform element 330 is furtherprocessed by a respective demodulator, 335-i, at each of the transpondercarrier frequencies. An illustrative demodulator is shown in FIG. 17 forprocessing an output signal from transform element 330 into a basebandIQ signal for a respective transponder channel. In this illustrativeembodiment, one demodulator of the form illustrated in FIG. 17 isrequired for each desired transponder channel.

As described above, receiver 200 enables a plurality of frequencychannels to be simultaneously tuned such that broadcast channel programsincluded within different frequency channels may be simultaneouslyaccessed. In addition, and in accordance with an aspect of theinvention, the amount of hardware and processing required to implement amulti-channel tuner is simplified by use of a single computation elementas represented by transform element 330. For example, now allcalculations are performed at a convenient rate, e.g., F_(F).

As noted above, the transform element may be implemented in anintegrated circuit such as an FPGA. As such, as shown in FIG. 18, asingle-chip solution is possible for simultaneously providing contentfrom different transponder channels. Illustratively, an integratedcircuit 400 may include at least a transform element, such asrepresented by transform element 330, described above, to providetherefrom a plurality of virtual channels 240-1 to 240-K, where at leastsome of the content of these virtual channels are simultaneouslyprovided from different transponder channels. As required, theintegrated circuit 400 may include demodulators (DM), as describedabove.

It should be noted that other forms of LNB processing may also be used.For example, LNB 205 may perform a filtering operation to a relaxedspecification with a broad transition band of width (PF_(S)) above andbelow the N channel band to reach acceptable stop band attenuation,where P is an integer. Moreover, the LNB may spectrally move the lowestfrequency channel so that the corresponding carrier F₀ is equal to[F_(S)/2+(PF_(S))]. With this variation, the A/D converter 210 isclocked at the sampling rate [2(N+(2P))F_(S)], and the number ofdemultiplexer parallel paths used for signal tuning is N+(2P). Thisvariation may allow LNB 205 to utilize smaller, lower performancefilters, rather than physically larger and lossy SAW filters.

Similarly, LNB 205 may provide signal 206 such that the frequency of thehighest frequency channel (i.e., F_(N)) is arranged to fall on an evenfolding frequency of the demultiplexer sampling rate, F_(F). Thistechnique may be used for those highest frequency channels that satisfy:

$\begin{matrix}{{F_{F} = {2\left\lbrack \frac{F_{N} + \frac{F_{S}}{2}}{2N\; F_{S}} \right\rbrack}},} & (7)\end{matrix}$when sampling A/D 210 at 2NF_(S), or

$\begin{matrix}{{F_{F} = \left\lbrack \frac{F_{N} + {F_{S}\left( {P + {.5}} \right)}}{2\left( {N + {2P}} \right)F_{S}} \right\rbrack},} & (8)\end{matrix}$when sampling A/D 210 at [2 (N+(2P))F_(S)].

Likewise, LNB 205 may provide signal 206 such that the frequency of thelowest frequency channel (i.e., F₁) is arranged to fall on an evenfolding frequency of the demultiplexer sampling rate, F_(F). Thistechnique may be used for those lowest frequency channels that satisfy:

$\begin{matrix}{{F_{F} = {2\left\lbrack \frac{\frac{F_{1} - F_{2}}{2}}{2N\; F_{S}} \right\rbrack}},} & (9)\end{matrix}$when sampling A/D 210 at 2NF_(S), or

$\begin{matrix}{{F_{F} = \left\lbrack \frac{F_{1} - {F_{S}\left( {P + {.5}} \right)}}{2\left( {N + {2P}} \right)F_{S}} \right\rbrack},} & (10)\end{matrix}$when sampling A/D 210 at [2 (N+(2P))F_(S)].

It should also be noted that constraints on the clock rate of A/D 210can be relaxed somewhat by inclusion of a sample rate converter. Thelatter representing a calculated sequence derived from some sampling(uniform or non-uniform) not conforming to the desired sample spacing T.In addition, it should be noted that other types of DCTs may be also beused in accordance with the inventive concept, e.g., type 2, type 3,etc. However, mismatches of boundary conditions in time and frequencyentail a hardware and circuit complexity penalty for using these typesof DCTs and, as such, are not described further herein.

Further, it should be noted that although described in the context of asatellite distribution, the inventive concept is not so limited and alsoapplies to other distribution mechanisms whether wireless and/or wired.For example, the invention is applicable to cable, terrestrial or othernetworks (such as broadcast and/or commercial networks).

As such, the foregoing merely illustrates the principles of theinvention and it will thus be appreciated that those skilled in the artwill be able to devise numerous alternative arrangements which, althoughnot explicitly described herein, embody the principles of the inventionand are within its spirit and scope. For example, although illustratedin the context of separate functional elements, these functionalelements may be embodied on one or more integrated circuits (ICs).Similarly, although shown as a separate elements, any or all of theelements of FIGS. 10 and 12 (e.g., 215 and/or 240) may be implemented ina stored-program-controlled processor. It is therefore to be understoodthat numerous modifications may be made to the illustrative embodimentsand that other arrangements may be devised without departing from thespirit and scope of the present invention as defined by the appendedclaims.

1. A receiver comprising: a receiver section for providing a signalhaving a plurality of different frequency channels, each frequencychannel conveying a different bit stream; and a multi-channel signaltuner operative on the signal for recovering the different bit streamsfrom at least two of the plurality of different frequency channels andfor simultaneously providing the recovered different bits streams,wherein the multi-channel signal tuner utilizes a discrete cosinetransformation (DCT) wherein the multi-channel signal tuner comprises: asampler for sampling the signal to provide a number of decimated samplestreams; a transform element operative on the number of decimated samplestreams to provide transform output signals wherein the transformelement utilizes the DCT; and a number of demodulators operative on thetransform output signals to provide the recovered different bit streams.2. The receiver of claim 1, wherein the sampler includes: ademultiplexer for demultiplexing the signal into the number of decimatedsample streams; and a divided filter bank for processing the number ofdecimated sample streams for matching the DCT utilized by the transformelement.
 3. The receiver of claim 2, wherein the divided filter bankincludes a bank of bifurcated filters and the DCT is an N-point type IVDCT.
 4. The receiver of claim 3, wherein the transform element performsmatrix-based processing on the number of decimated sample steams using asparse matrix factorization of the N-point type IV DCT.
 5. The receiverof claim 1, further comprising a broadcast channel distributor forproviding a number of virtual channels from the recovered different bitstreams.
 6. A satellite receiver comprising: a low-noise block forreceiving a signal representing a plurality of different transponderchannels and an analog-to-digital converter for providing therefrom adata signal representing a sequence of samples occurring at a samplerate greater than, or equal to, a Nyquist rate related to a totalbandwidth of the plurality of different transponder channels, eachtransponder channel conveying a bit stream; a sampler for sampling thedata signal for providing N decimated data streams, where N>1; atransform element operative on the N decimated data streams forsimultaneously providing at least two transform output signals from atleast two of the plurality of different transponder channels, whereinthe transform element utilizes a discrete cosine transformation (DCT);and at least two demodulators operative on the at least two transformoutput signals to provide at least two bit streams from the plurality ofdifferent transponder channels.
 7. The satellite receiver of claim 6,wherein the sampler includes: a demultiplexer for demultiplexing thesignal into the number of decimated data streams; and a divided filterbank for processing the number of decimated sample streams for matchingthe DCT utilized by the transform element.
 8. The satellite receiver ofclaim 6, wherein the divided filter bank includes a bank of bifurcatedfilters and the DCT is an N-point type IV OCT.
 9. The satellite receiverof claim 8, wherein the transform element performs matrix-basedprocessing on the number of decimated data steams using sparse matrixfactorization.
 10. The satellite receiver of claim 6, further comprisinga broadcast channel distributor for providing a number of virtualchannels from the at least two bit streams.
 11. A multi-channel tunercomprising: a sampler for sampling a signal representing a plurality ofdifferent frequency channels to provide a plurality of sample streams,each frequency channel conveying a transport bit stream; a bank offilters for filtering the plurality of sample streams to provide aplurality of filtered sample streams; a discrete cosine based transformelement operative on the plurality of filtered sample streams to providea plurality of data signals, each data signal associated with one of thedifferent frequency channels; and a plurality of demodulators fordemodulating each of the plurality of data signals to simultaneouslyprovide the transport bit streams.
 12. The multi-channel tuner of claim11 wherein the plurality of data signals is at least two.
 13. Themulti-channel tuner of claim 11, wherein the bank of filters include aplurality of bifurcated filters and the discrete cosine based transformelement uses a type IV discrete cosine transform.
 14. A method for usein a receiver comprising: providing a signal having a plurality ofdifferent frequency channels, each frequency channel conveying adifferent bit stream; and performing multi-channel signal tuning on thesignal for recovering the different bit streams from at least two of theplurality of different frequency channels; and simultaneously providingthe recovered different bits streams, wherein the step of multi-channelsignal tuning utilizes a discrete cosine transformation (DCT) andwherein the multi-channel signal tuning step comprises; sampling thesignal to provide a number of decimated sample streams; performingtransform-based processing on the number of decimated sample streams toprovide transform output signals wherein the transform based processingutilizes the DCT; and demodulating the transform output signals toprovide the recovered different bit streams.
 15. The method claim 14,wherein the sampling step includes: demultiplexing the signal into thenumber of decimated sample streams; and processing the number ofdecimated sample streams with a divided filter bank for matching the DCTutilized by the transform element.
 16. The method of claim 15, whereinthe divided filter bank includes a bank of bifurcated filters and theDCT is an N-point type IV DCT.
 17. The method of claim 14, wherein theperforming transform-based processing step performs matrix-basedprocessing on the number of decimated sample steams using a sparsematrix factorization of the N-point type IV DCT.
 18. The method of claim14, further comprising the step of providing a number of virtualchannels from the recovered different bit streams.